1. Field of the Invention
This present invention generally relates to the field of inductors, and more particularly, to an inductor with variable inductance.
2. Description of the Prior Art
Some cost-effective power converters with power factor correction (PFC) for universal-line-voltage (90-270 Vrms) applications require a variable PFC inductance to meet the requirements for line-current harmonics and power factor set by different standards and programs. For example, Light-Emitting Diode (LED) drivers with an input power over 25 W in general lighting applications are required to meet the line-current-harmonic limits set by the IEC 61000-3-2 Class C and JIS C 61000-3-2 Class C standards.
A good candidate for the universal-line LED driver applications is the single-stage PFC flyback topology shown in FIG. 1, as disclosed in U.S. Pat. No. 6,950,319 to L. Huber, M. M. Jovanović, and C. C. Chang, entitled “AC/DC flyback converter,” due to its low component count and low cost. In this converter, the PFC part operates in discontinuous conduction mode (DCM), while the dc/dc part operates at the DCM/CCM (continuous conduction mode) boundary. A low line-current harmonic distortion can be achieved due to the inherent property of the DCM boost converter to draw a near sinusoidal current if its duty cycle is held relatively constant during a half line cycle. However, voltage VB across bulk capacitor CB is not regulated and at high line it can increase to impractical levels. To reduce the bulk-capacitor voltage, one terminal of the boost inductor winding is connected to a tapping point of the primary winding of the flyback transformer, which provides a negative magnetic feedback. However, the tapping of the flyback primary winding also results in a zero-crossing distortion of the line current. In fact, as long as the instantaneous line voltage is lower than the voltage at the tapping point, no current is drawn from the input, which deteriorates the power factor and the line-current harmonics.
The single-stage PFC flyback topology with a constant inductance LB in FIG. 1 has been successfully applied in adapter/charger applications for the universal line voltage, where the line-current harmonics have to meet the IEC 61000-3-2 Class D and JIS C 61000-3-2 Class D limits. However, applying the single-stage PFC flyback topology with a constant inductance LB in FIG. 1 for lighting applications, where the line-current harmonics have to meet the more stringent limits set by the IEC 61000-3-2 Class C and JIS C 61000-3-2 Class C standards, presents a challenging task.
As voltage VB across bulk-capacitor CB in FIG. 1 is not regulated and varies with both the input voltage and output power, the design of the magnetic components significantly affects the bulk-capacitor voltage level. Generally, a higher boost inductance LB leads to a lower voltage VB. In fact, if the boost inductance increases during steady-state operation, the input power initially decreases because of a lower input current. The difference between the output power and input power has to be supplied from the bulk capacitor, causing a drop of the bulk-capacitor voltage. Meanwhile, as the bulk-capacitor voltage decreases, the duty cycle of main switch Q1 increases to keep the output voltage regulated, resulting in an increase of the input power until a new balance between the input and output power is reached. A higher boost inductance can limit voltage VB to an acceptable level (i.e., less than 450 V) and ensure DCM operation at high line (180-270 Vrms). However, at low line (90-135 Vrms), if the boost inductance is larger than the maximum value for DCM operation, the boost inductor will enter CCM operation around the peak of the rectified line voltage, and the line current waveform will have a bulge around its peak value, resulting in an increased total harmonic distortion (THD). Furthermore, if the bulk-capacitor voltage is slightly lower than the peak of the rectified line voltage, peak charging of the bulk capacitor through the bridge rectifier will also result in a bulge in the line current waveform with an increased THD.
It was shown in “Single-stage flyback power-factor-correction front-end for high-brightness (HB) LED application,” by Y. Hu, L. Huber, and M. M. Jovanović, Proc. IEEE Industry Applications Society (IAS) 2009, that the single-stage PFC flyback in FIG. 1 with a constant boost inductance cannot be designed to achieve a practical bulk-capacitor voltage level at high line while meeting the JIS C 61000-3-2 Class C line-current harmonic limits at low line. To overcome these limitations, a variable boost inductance is required, i.e., a high boost inductance at high line to limit the bulk-capacitor voltage and a lower boost inductance at low line to ensure DCM operation and a low THD.
Inductors with variable inductance are known in prior art and they can be classified in three groups.
The first group includes methods where the inductance is varied by changing the path of the magnetic flux by using a short-circuited control winding. For example, see 1) U.S. Pat. No. 3,873,910 to C. A. Willis, entitled “Ballast control device,” and 2) U.S. Pat. No. 4,162,428 to Robert T. Elms, entitled “Variable inductance ballast apparatus for HID lamp.”
FIG. 2 shows a prior art variable inductor for use in lamp ballasts, as disclosed in U.S. Pat. No. 3,873,910. As shown in FIG. 2, the variable inductor comprises a main winding and a control winding positioned on the opposite sides of an added, gapped shunt. When the control winding is shorted by closing the triac switch, a current flows through the control winding generating a magnetic flux opposing the main flux induced by the main winding. As a result, the main flux path is forced to pass through the shunt. Since the gapped magnetic shunt has a higher reluctance than the flux path around the closed core, the inductance of the device is decreased. Therefore, the lamp current and the lamp power are increased.
FIG. 3 shows a prior art variable inductor for use in ballasts for HID lamps, as disclosed in U.S. Pat. No. 4,162,428. The control winding is wound around one outer leg of the EI magnetic core. Similarly as in the previous case, when the control switch is closed, a current flows through the control winding generating a magnetic flux opposing the main flux induced by the main winding, and the main flux path is forced to pass through the other outer leg of the core which has a gap, causing a decrease of the inductance and increase of the lamp current and power.
A major drawback of the methods disclosed in U.S. Pat. No. 3,873,910 and No. 4,162,428 is that a short circuit is created when the control switch is closed, resulting in a significant power loss in the control winding and switch.
In the second group, the inductance is varied by changing the size of the non-magnetic gap along the magnetic flux path either mechanically by using, for example, an actuator made of piezoceramic material that changes its length in response to an applied voltage, as disclosed in U.S. Pat. No. 5,999,077 to R. E. Hammond, E. F. Rynne, and L. J. Johnson, entitled “Voltage controlled variable inductor,” or by a non-uniform gap construction such as a stepped gap or a sloped gap as described in “Quasi-active power factor correction with a variable inductive filter: theory, design and practice” by W. H. Wölfle and W. G. Hurley, IEEE Transactions on Power Electronics, vol. 18, no. 1, pp. 248-255, January 2003.
In U.S. Pat. No. 5,999,077, a voltage-controlled variable inductance is disclosed, where an actuator, made of piezoceramic material that changes its length in response to an applied voltage, is fastened in the window area of the core in order to change the length of the air gap between the two parts of the magnetic core, resulting in a variation of the inductance. However, the inclusion of the actuator requires a complex implementation.
In the paper by Wölfle, variation of the inductance is achieved by varying the length of the air gap either in a discrete step (stepped air gap) or with a graded slope (sloped air gap). The value of the inductance varies with the inductor current. In fact, the core of the inductor with the stepped air gap (also called swinging inductor) can be considered to have two parallel reluctance paths, each path having two reluctances in series, the core and the gap. As the current increases, the path containing the smaller gap reaches saturation first and the increased reluctance reduces the overall inductance. The sloped air-gap inductor operates on the same principle; however, the variation of the inductance with the current is more gradual. Generally, manufacturing inductors with a stepped or sloped air gap is more complex than manufacturing inductors with a constant-length air gap, resulting in an increased cost.
The variable inductors built by using powdered metal cores with distributed air gap (see, for example, www.mag-inc.com/products/powder_cores) also belong to the second group. The powdered metal cores exhibit a soft saturation property, i.e., their permeability gradually decreases as the magnetizing force increases. However, the powdered metal cores have significantly higher loss than the corresponding ferrite cores.
The third group includes methods where the inductance is varied by adding a dc bias flux to the main magnetic flux. For example, see 1) U.S. Pat. No. 4,992,919 to C. Q. Lee, K. Siri, and A. K. Upadhyay, entitled “Parallel resonant converter with zero voltage switching;” 2) “Quasi-linear controllable inductor” by A. S. Kislovski, Proceedings of the IEEE, vol. 75, no. 2, pp. 267-269, February 1987, (Kislovski, 1987); 3) U.S. Pat. No. 4,853,611 to A. Kislovski, entitled “Inductive, electrically-controllable component;” 4) “Relative incremental permeability of soft ferrites as a function of the magnetic field H: an analytic approximation,” by A. S. Kislovski, Rec. IEEE Power Electronics Specialists Conf. (PESC), pp. 1469-1475, 1996, (Kislovski, 1996); 5) “A current-controlled variable-inductor for high frequency resonant power circuits” by D. Medini and S. B. Yaakov, Proc. IEEE Applied Power Electronics Conf. (APEC), pp. 219-225, 1994; and 6) U.S. Pat. No. 4,393,157 to G. Roberge and A. Doyon, entitled “Variable inductor.”
FIG. 4 shows a prior art variable inductor where an inductor winding and a control winding are wound on the same magnetic core, as disclosed in U.S. Pat. No. 4,992,919. A dc bias current IBIAS flowing through the control winding produces a bias magnetic flux ΦC. The main magnetic flux ΦLO produced by the inductor current is superimposed on the bias magnetic flux ΦC.
FIG. 5 shows a graph of the relationship between the magnetizing field (H) and magnetic field (B) for the prior art variable inductor of FIG. 4. As the dc bias magnetizing force increases, the permeability of the core material, i.e., the slope of the B-H curve
  (      =                  lim                              Δ            ⁢                                                  ⁢            H                    ->          0                    ⁢              Δ        ⁢                                  ⁢                  B          /          Δ                ⁢                                  ⁢        H              )decreases, leading to a decreased inductance. A drawback of this method is that the control winding is strongly coupled with the inductor winding, resulting in undesired induced ac current and, consequently, power loss in the control winding.
FIG. 6 shows a prior art variable inductor where the inductor winding is divided into two identical portions, which are wound on two identical toroidal cores and connected in series so as to produce opposing fluxes through the control winding, which is wound over both cores, as proposed in the paper by Kislovski, 1987, and in U.S. Pat. No. 4,853,611. Ideally, due to the opposing fluxes, there is no coupling between the inductor and control windings.
FIG. 7 shows graphs of the relationship between the magnetizing field (H) and magnetic field (B) for the two individual cores of the variable inductor in FIG. 6: (a) without and (b) with a control current in the control winding. As shown in FIG. 7(a), without a bias current in the control winding, both cores exhibit the same flux density variation, i.e., ΔB1=ΔB2, Therefore, the total inductance is twice the inductance of the individual inductor windings. Additionally, the induced voltage in the control winding is zero due to the equal but opposing fluxes through the control winding. However, with a bias current in the control winding, a biasing field H0 is produced which displaces the operating point of the cores along their B-H curves, as shown in FIG. 7(b). One core (core 1) operates in the non-linear to saturation region, whereas the other core (core 2) operates in the non-linear to linear region along their respective B-H curves. As a result, the flux density variation and, consequently, the permeability in both cores are reduced compared to the case without a DC bias. Therefore, the total inductance is reduced. In addition, the flux density variation in core 1 is smaller than that in core 2, i.e., ΔB1<ΔB2. Consequently, the total flux density variation through the control winding is not zero, resulting in undesired induced ac voltage and power loss in the control winding.
FIG. 8 shows another prior art implementation of the variable inductor in FIG. 6 as described in the paper by Kislovski, 1996, where instead of two toroidal cores a pair of E cores is used.
FIG. 9 shows a prior art modification of the variable inductor in FIG. 6 as disclosed in U.S. Pat. No. 4,853,611, where both the inductor winding and control winding are each divided into two identical portions, wound on two toroidal cores, and connected in series. Under this arrangement, the principle of operation and, consequently, the drawbacks of the variable inductor in FIG. 9 are the same as those of the variable inductor in FIG. 6.
FIG. 10 shows a prior art variable inductor, as proposed in the paper by Medini, which is similar to the variable inductor in FIG. 8, with the difference that the positions of the inductor winding and control winding are flipped. Specifically, the control winding is divided into two identical portions wound around the outer legs of the EE core and connected in series, while the inductor winding is wound around the center leg of the EE core. Also, the air gaps from the outer legs of the EE core are moved to the central leg. Under this arrangement, the basic operation and, consequently, the drawbacks of the variable inductor in FIG. 10 are the same as those of the variable inductor in FIG. 6.
In U.S. Pat. No. 4,393,157, a dc bias flux is added orthogonally to the main magnetic flux, which requires a complex magnetic core structure. In addition, orthogonal-flux inductors exhibit a smaller inductance variation than the parallel-flux inductors at the same control-current variation, as explained in “Comparison of orthogonal- and parallel-flux variable inductors,” by Z. H. Meiksin, IEEE Trans. Industry Applications, vol. IA-10, no. 3, pp. 417-423, May/June 1974.
The drawback of all current-controlled variable inductors in FIGS. 4, 6, and 8-10 is that the control winding is always coupled to the inductor winding. Even with two opposing fluxes through the control winding produced by the inductor winding, there is always an asymmetry in the operation, such as shown in FIG. 7(b). Therefore the opposing fluxes do not completely cancel each other, resulting in undesired induced ac voltage and, consequently, increased power loss in the control winding. In addition, any asymmetry in the structure of the magnetic core and any mismatch in the two portions of the inductor winding or control winding further increase the unbalance between the opposing fluxes through the control winding and increase the undesired induced ac voltage and power loss in the control winding. In ripple-sensitive applications such as LED drivers, any additional ripple in the LED current would adversely affect the longevity of the LEDs.
Therefore, there exists a need for an inductor that provides a variable inductance with a simple control technique without significantly affecting efficiency and without significantly affecting load current.